Inverters pulse width modulation techniques, Thesis for Power Electronics

Inverters pulse width modulation techniques, Thesis for Power Electronics

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2.1 Introduction

The dc-ac converter, also known as the inverter, converts dc power to ac power

at desired output voltage and frequency. The dc power input to the inverter is

obtained from an existing power supply network or from a rotating alternator through

a rectifier or a battery, fuel cell, photovoltaic array or magneto hydrodynamic

generator. The filter capacitor across the input terminals of the inverter provides a

constant dc link voltage. The inverter therefore is an adjustable-frequency voltage

source. The configuration of ac to dc converter and dc to ac inverter is called a dc-

link converter.

Inverters can be broadly classified into two types, voltage source and current

source inverters. A voltage–fed inverter (VFI) or more generally a voltage–source

inverter (VSI) is one in which the dc source has small or negligible impedance. The

voltage at the input terminals is constant. A current–source inverter (CSI) is fed with

adjustable current from the dc source of high impedance that is from a constant dc


A voltage source inverter employing thyristors as switches, some type of forced

commutation is required, while the VSIs made up of using GTOs, power transistors,

power MOSFETs or IGBTs, self commutation with base or gate drive signals for their

controlled turn-on and turn-off.


A standard single-phase voltage or current source inverter can be in the half-

bridge or full-bridge configuration. The single-phase units can be joined to have

three-phase or multiphase topologies. Some industrial applications of inverters are for

adjustable-speed ac drives, induction heating, standby aircraft power supplies, UPS

(uninterruptible power supplies) for computers, HVDC transmission lines, etc.

In this chapter single-phase inverters and their operating principles are

analyzed in detail. The concept of Pulse Width Modulation (PWM) for inverters is

described with analyses extended to different kinds of PWM strategies. Finally the

simulation results for a single-phase inverter using the PWM strategies described are


2.2 Voltage Control in Single - Phase Inverters

The schematic of inverter system is as shown in Figure 2.1, in which the

battery or rectifier provides the dc supply to the inverter. The inverter is used to

control the fundamental voltage magnitude and the frequency of the ac output

voltage. AC loads may require constant or adjustable voltage at their input terminals,

when such loads are fed by inverters, it is essential that the output voltage of the

inverters is so controlled as to fulfill the requirement of the loads. For example if the

inverter supplies power to a magnetic circuit, such as a induction motor, the voltage

to frequency ratio at the inverter output terminals must be kept constant. This avoids

saturation in the magnetic circuit of the device fed by the inverter.


Battery or

Rectifier Inverter




Figure 2.1: Schematic for Inverter System

The various methods for the control of output voltage of inverters can be classified as:

(a) External control of ac output voltage

(b) External control of dc input voltage

(c ) Internal control of the inverter.

The first two methods require the use of peripheral components whereas the third

method requires no external components. Mostly the internal control of the inverters

is dealt, and so the third method of control is discussed in great detail in the following


2.2.1 Pulse Width Modulation Control

The fundamental magnitude of the output voltage from an inverter can be

controlled to be constant by exercising control within the inverter itself that is no

external control circuitry is required. The most efficient method of doing this is by

Pulse Width Modulation (PWM) control used within the inverter. In this scheme the

inverter is fed by a fixed input voltage and a controlled ac voltage is obtained by


adjusting the on and the off periods of the inverter components. The advantages of the

PWM control scheme are [10]:

a) The output voltage control can be obtained without addition of any external


b) PWM minimizes the lower order harmonics, while the higher order

harmonics can be eliminated using a filter.

The disadvantage possessed by this scheme is that the switching devices used in the

inverter are expensive as they must possess low turn on and turn off times,

nevertheless PWM operated are very popular in all industrial equipments. PWM

techniques are characterized by constant amplitude pulses with different duty cycles

for each period. The width of these pulses are modulated to obtain inverter output

voltage control and to reduce its harmonic content. There are different PWM

techniques which essentially differ in the harmonic content of their respective output

voltages, thus the choice of a particular PWM technique depends on the permissible

harmonic content in the inverter output voltage.

2.2.2 Sinusoidal-Pulse Width Modulation (SPWM)

The sinusoidal PWM (SPWM) method also known as the triangulation, sub

harmonic, or suboscillation method, is very popular in industrial applications and is

extensively reviewed in the literature [1-2]. The SPWM is explained with reference to

Figure 2.2, which is the half-bridge circuit topology for a single-phase inverter.




dV 2 dV

2 dV






Figure 2.2: Schematic diagram for Half-Bridge PWM inverter.

For realizing SPWM, a high-frequency triangular carrier wave is

compared with a sinusoidal reference of the desired frequency. The intersection of

and waves determines the switching instants and commutation of the modulated

pulse. The PWM scheme is illustrated in Figure 2.3 a, in which v is the peak value of

triangular carrier wave and v that of the reference, or modulating signal. The figure

shows the triangle and modulation signal with some arbitrary frequency and

magnitude. In the inverter of Figure 2.2 the switches and are controlled based

on the comparison of control signal and the triangular wave which are mixed in a

comparator. When sinusoidal wave has magnitude higher than the triangular wave the

comparator output is high, otherwise it is low.



cv rv




11S S

v > is on , r cv 11S 2 d

out V =V (2.1)


< is on , rv cv 12S 2 d

out V −=V (2.2)




Figure 2.3: SPWM illustration (a) Sine-Triangle Comparison (b) Switching Pulses after comparison.


The comparator output is processesed in a trigger pulse generator in such a

manner that the output voltage wave of the inverter has a pulse width in agreement

with the comparator output pulse width. The magnitude ratio of c


v v is called the

modulation index ( ) and it controls the harmonic content of the output voltage

waveform. The magnitude of fundamental component of output voltage is

proportional to . The amplitude of the triangular wave is generally kept

constant. The frequency–modulation ratio is defined as


im cv



t f f

f m = (2.3)

To satisfy the Kirchoff’s Voltage law (KVL) constraint, the switches on the same leg

are not turned on at the same time, which gives the condition

+ = 1 (2.4) 11S 12S

for each leg of the inverter. This enables the output voltage to fluctuate between

2 dV and

2 dV− as shown in Figure 2.4 for a dc voltage of 200 V.


Figure 2.4: Output voltage of the Half-Bridge inverter.

2.3 Single-Phase Inverters

A single-phase inverter in the full bridge topology is as shown in Figure

2.5, which consists of four switching devices, two of them on each leg. The full-

bridge inverter can produce an output power twice that of the half-bridge inverter

with the same input voltage. Three different PWM switching schemes are discussed

in this section, which improve the characteristics of the inverter. The objective is to

add a zero sequence voltage to the modulation signals in such a way to ensure the

clamping of the devices to either the positive or negative dc rail; in the process of

which the voltage gain is improved, leading to an increased load fundamental voltage,

reduction in total current distortion and increased load power factor. In Figure 2.5, the

top devices are assigned to be S11 and S21 while the bottom devices as S12 and S22, the

voltage equations for this converter are as given in the following equations.




dV abV





2 dV

2 dV





o +


Figure 2.5: Schematic of a Single Phase Full-Bridge Inverter.

aonoan d VVVSSV =+=− )(

2 1211 (2.5)

bonobn d VVVSSV =+=− )(

2 2221 (2.6)

bnanab VVV −= (2.7)

The voltages and V are the output voltages from phases A and B to an

arbitrary point n, V is the neutral voltage between point n and the mid-point of the

DC source. The switching function of the devices can be approximated by the Fourier

series to be equal to

anV bn


)M1( + 2 1 where M is the modulation signal which when

compared with the triangular waveform yields the switching pulses [19]. Thus from

Equations 2.4, 2.5, and 2.6, the expressions for the modulation signals are obtained as




M )(2

11 +

= (2.8)

. )(2

21 d



M +

= (2.9)


Equations 2.8 and 2.9 give the general expression for the modulation signals for

single-phase dc-ac converters. The various types of modulation schemes presented in

the literature can be obtained from these equations using appropriate definition for

, V and V . Making use of this concept different modulation schemes have been

proposed some of which are explained in detail in the following sections.

anV bn no

2.3.1 SPWM With Bipolar Switching

In this scheme the diagonally opposite transistors S11, S22 and S21 and S12 are

turned on or turned off at the same time. The output of leg A is equal and opposite to

the output of leg B. The output voltage is determined by comparing the control signal,

and the triangular signal, V as shown in Figure 2.6(a) to get the switching pulses

for the devices , and the switching pattern is as follows.

rV c

rV >V , Sc 11 is on => 2 Vd

ao =V and S22 is on => 2 Vd

bo −=V ; (2.10)

rV <V , Sc 12 is on => 2 Vd

ao −=V and S21 is on => 2 Vd

bo =V ; (2.11)


)()( tVtV aobo −= (2.12)





Figure 2.6:Bipolar PWM (a) Sine-triangle comparison (b) Switching pulses for S11/S22 (c) Switching pulses for S12/S21




Figure 2.7: Bipolar PWM scheme (a) Modulation signal for leg ‘a’ (b) output line-line voltage (c) load current


The line-to-line voltage is given as in Equation 2.13.

)V (2.13) (2)()()( tVtVtVt aoboaoab =−=

The peak of the fundamental-frequency component in the output voltage is given as


diab VmV = ( ) (2.14) 0.1≤im


dabd VVV π 4 << (m ). (2.15) 0.1≥i

Since the voltage switches between two levels dV− and V , the scheme is called the

Bipolar PWM. The relationship between fundamental input and output voltage in the

overmodulating region is given as [10].


do MVV = (2.16)


)1(sin2 21 ααα π

−+= −i mM ,m 1>i

im/1=α .

For a full-bridge inverter with bipolar PWM scheme the output voltage is between

2 dV− and

2 dV . Figure 2.7 shows the modulation signal, output voltage, and the load

current for bipolar modulation scheme on a single-phase inverter with an RL load of

10 Ω and 0.125H.


For the bipolar PWM switching scheme there is only one modulation signal and the

switches are turned ‘on’ or turned ‘off’ according to the pattern given in Equations

2.10 and 2.11. The input dc voltage was 200 V and the modulation index (mi) was

taken to be 0.8. The switching frequency for the carrier, which is the triangle, is 10


2.3.2 SPWM With Unipolar Switching

In this scheme, the devices in one leg are turned on or off based on the

comparison of the modulation signal V with a high frequency triangular wave. The

devices in the other leg are turned on or off by the comparison of the modulation

signal with the same high frequency triangular wave. Figure 2.8 and 2.9 show

the unipolar scheme for a single –phase full bridge inverter, with the modulation

signals for both legs and the associated comparison to yield switching pulses for both

the legs.



In Figure 2.8 the simulation results show the sine triangle comparison, the

switching pulses for S11 and S21 are shown. The switching for the other two devices is

obtained as S12 = 1 – S11 and S22 = 1- S21. Figure 2.9 shows the phase voltages , line-

to-line voltages obtained from a unipolar PWM scheme , also shown is the load

current. The simulation was carried out for an RL load of R = 10Ω and L = 0.125H.

The dc voltage is 200 V and the switching frequency is 10kHz. The modulation signal

has a magnitude of 0.8, i.e mi = 0.8.





Figure 2.8: Unipolar PWM voltage switching scheme (a) Sine triangle comparison (b) switching pulses for S11 (c) switching pulses for S21.





Figure 2.9: Unipolar PWM voltage switching scheme (a) phase voltage ‘a’ (b) phase voltage ‘b’ (c) line to line voltage Vab (d) load current


The logic behind the switching of the devices in the leg connected to ‘a’ is

given as

rV > V : is on and V = c 11S an 2 dV (2.17)

rV <V : is on and V = c 11S an 2 dV− (2.18)

and that in the leg connected to ‘b’ is given as

-V >V : is on and V = r c 11S bn 2 dV (2.19)

-V <V : is on and V = r c 11S bn 2 dV− (2.20)

Table 2.1 shows the switching state of the unipolar PWM and the

corresponding voltage levels. It can be observed from the table that when the two top

or the two bottom devices are turned on the output voltage is zero.

In Unipolar switching scheme the output voltage level changes between

either 0 to -V or from 0 to +V . This scheme ‘effectively’ has the effect of doubling

the switching frequency as far as the output harmonics are concerned, compared to

the bipolar- switching scheme. The voltage waveforms V and are 180o out of

phase from each other as seen in Figure 2.10. The output voltage V is as shown in

Figure 2.11 along with the load current.

d d

an bnV


Since the harmonic components at the switching frequency in and

have the same frequency, this results in the cancellation of the harmonic

component at the switching frequency in the output voltage.




Table 2.1. Switching state of the unipolar PWM and the corresponding voltage levels.

11S 12S 21S 22S AnV BnV BnAno VVV −=

ON - - ON dV 0 dV

- ON ON - 0 dV -V d

ON - ON - dV dV 0

- ON - ON 0 0 0

The fundamental component of the output voltage is given as

dio VmV = ( ) (2.21) 0.1≤im

dod VVV π 4 << (m ). (2.22) 0.1>i

2.3.3 SPWM With Modified Bipolar Switching Scheme (MBPWM)[14]

In the inverter employing the bipolar switching scheme, switches are

operated in such a way that during the positive half of the modulation signal one of

the top devices in one of the switching leg is kept on and the two other switching

devices in the other leg are PWM operated, and during the negative half of the

modulation signal one of the bottom switching device is kept on continuously while

the other two switching devices in the other leg are PWM operated. The output

voltage is determined by comparing the control signal and the triangular wave. rV


The switching pattern along with the sine-triangle comparison is as shown in Figure

2.10. The switching pattern for positive values of modulating signal V is as given m

V > V , is on (2.23) r c 21S

and V <V , is on . r c 22S






Figure 2.10: Modified bipolar PWM (a) Sine-triangle comparison (b), (c), (d), and (e) switching pulses for devices S11, S12, S21 and S22.




Figure 2.11: Modified bipolar PWM scheme (a) line-to-line voltage (b) load current

The switching pattern for negative values of the modulating signal V is given as m

V < V , is on (2.24) r c 21S

and V > V , is on . r c 22S

The output voltage is given as V )()()( tVtVt BnAno −= , as shown in Figure 2.11. The

load current is also shown in the same plot. The RL load has an R = 10 Ω and L =

0.125H. The modulation signal for the sine-triangle comparison is 0.8. The switching

pattern for the Modified Bipolar Switching Scheme is as given in Table 2.2.


Table 2.2. Switching state of the modified bipolar PWM and the corresponding voltage.

11S 12S 21S 22S AnV BnV BnAno VVV −=

ON - - ON dV 0 dV

- ON ON - 0 dV -V d

ON - ON - dV dV 0

- ON - ON 0 0 0

From Table 2.2 it can be observed that when the two top or the two bottom devices

are turned on the output voltage is zero.

In the modified bipolar switching scheme the output voltage level changes

between either 0 to -V or from 0 to +V . Since the sign of the modulation signal

decides the switching pattern the analysis of this switching scheme is complex. The

relationship between input and output voltage is given as [14],

d d

do mVV =


where )4(5.0 π += imm ( 0.1<im ) . (2.26)

Thus from the above equation it can be observed that the fundamental component of

the voltage as obtained from the MBPWM is the maximum when compared to the

other switching schemes even in the linear modulation region; that is when the

modulation index is less than unity.


2.3.4 Generalized Carrier-based PWM

In the inverter shown in Figure 2.5, the output voltage and the input current are

given as

noanaod VVVSSV +==− )(5.0 1211 (2.27)

nobnbod VVVSSV +==− )(5.0 2221 (2.28)

)( 2111 SSII ad −= (2.29)

bnanab VVV −= . (2.30)

The voltages V and V are the output voltages from phases ‘a’ and ‘b’ to a arbitrary

point while V is the neutral voltage between the point ‘n’ and the mid-point of the

DC source. The generalized carrier-based PWM scheme is obtained by defining the

quantity using the concept of q-d Space Vector representation. A special q-d

reference frame transformation to transform the two phase voltages to orthogonal q-d

voltage components is defined as





)(5.0 bnanq VVV += (2.31)

)(5.0 bnand VVV −= (2.32)

where and are the q-axis and the d-axis voltages in an orthogonal coordinate

system. The q-d voltages for each of the possible switching instant are shown in

Table 2.3.

qV dV


Table 2.3. Switching state of the generalized carrier based PWM scheme.

11S 21S aoV boV qV dV

- - dV5.0− dV5.0− nod VV −− 5.0 0

- ON dV5.0− dV5.0 noVdV5.0−

ON - dV5.0 dV5.0− noVdV5.0

ON ON dV5.0 dV5.0 nod VV −5.0 0

Figure 2.12 also shows the space vector representation of the output phase

voltages. To synthesize a given reference output voltage V or equivalently V , the

four vectors shown in the figure are averaged over one switching period for the


ab *


qdddqdccqdbbqdaaqd VtVtVtVtV +++= * (2.33)

where are the normalized times for which the averaging vector spent in

each of the four quadrants. The normalized times should satisfy the condition that

. The normalized times t can be expressed as some equivalent

time t

dcba tttt ,,,

1=++ dc tt+ ba tt dc t,

o such that

odc ttt =+ (2.34)

or equivalently t can be written as tdc t, oc tγ= which implies t od t)1( γ−= , ]10[∈γ so

Equation 2.33 can be written as

qddoqdcoqdbbqdaaqd VtVtVtVtV )1( * γγ −+++= (2.35)


qdb d fVVno =−− ]0, 2


qdc d fVVno =− ] 2


qda d fVnoV =− ]0, 2


qdd d fVVno =−− ] 2


* qdf

Figure 2.12: Space vector representation of the voltages in a single-phase inverter.

The time is the actual time which the vector spends in the null state that is

when either both the top or both the bottom devices are off or on at the same time.

This time is split in to two time periods such that


1=++ oba ttt ; let t then xba tt =+

xa tt ξ= and so xb tt )1( ξ−= , where ]1,0[∈ξ and ]1,0[∈γ . The quantities t re

the normalized times (with respect to the switching period of the converter). Solving

Equation 2.33 we can get the expression for the zero sequence voltage V in terms of

other known quantities as

oba tt ,


, a



2)12( )12(

)12(5.0 dd

dq dno VV



−− −−=

ξ ξ

γ (2.34)

Equations 2.8, 2.9, along with 2.34 constitute the generalized discontinuous PWM

scheme for the single-phase inverter. An infinite number of possibilities for the

discontinuous PWM exist depending on the choice of ξ and γ .


2.4 Bipolar and Modified Bipolar PWM Schemes with Zero Sequence Voltage

In the PWM modulation scheme with bipolar voltage switching, the

diagonally opposite switching devices are switched as switch pairs resulting in an

output voltage switching between -V and . The zero sequence voltage expression

for the bipolar schemes is given as,

d dV

5.0 )12( −= γdno VV as the q-axis voltage is zero

(refer Equations 2.31 and 2.32). If γ is so chosen so as to locate the zero sequence

voltage to be centered about the peak of the modulation signal, we can achieve higher

fundamental component of the load voltage and less switching because the effect of

the zero sequence is to increase the modulation signal to more than unity. In which

case the comparison of the triangle and the modulation signal would yield continuous

‘on’ or ‘off’ of the switching device for a long period of time as when compared to

the regular sine triangle comparison.

2.5 Implementation of the Bipolar and the Modified bipolar PWM Schemes for

an RL load

The single-phase inverter in the full-bridge topology has been simulated in

Matlab/Simulink for a RL load with R = 10Ω and L = 0.05 H. The modulation signals

(for a modulation index of 0.8) for the switching devices have been obtained from the

TMSLF2407, Texas Instruments DSP. Figure 2.13 shows the simulation result of

bipolar PWM with the zero-sequence voltage while without the zero sequence was

already shown in Figures 2.8 and 2.9. The simulation results for the modified bipolar

PWM scheme with the zero-sequence voltage are as shown in Figures 2.14. In the


simulation the dc voltage was assumed to be 200 V and the modulation index to be







Figure 2.13: Bipolar PWM scheme (a) modulation signal (b) &(c) switching pulses S11/S22 and S21/S12 respectively.









Figure 2.14: Modified bipolar PWM scheme (a) modulation signal (b), (c), (d) & (e) switching pulses S11, S12, S21 and S22 (f) line-to-line voltage (g) load current.


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