2001 Future Energy Challenge
Texas A&M University
Fuel Cell Inverter
- 10 kW Design & Cost Analysis
Texas A&M University, College Station, Texas
August 27, 2001
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2001 Future Energy Challenge
Texas A&M University
Fuel Cell Inverter
- 10 kW Design & Cost Analysis
Texas A&M University, College Station, Texas
August 27, 2001
TABLE OF CONTENTS: Page
frequency MOSFET and IGBT switches have been employed to achieve a higher efficiency, lower size and volume of the fuel cell inverter system.
DC-DC converter and inverter topologies were designed to achieve ease of manufacturability and mass production. Another unique aspect of the design is the use of the TMS320C2407 DSP to control the inverter. The DSP reduces printed circuit board layout complexity. Readily programmable, the DSP adds flexibility and intelligence to implement various control aspects by means of software. (See Appendix D for DSP cost information)
Two sets of lead-acid batteries are provided on the 200V DC bus to supply sudden load demands. By floating the standby battery off the 400V instead of at the 48V level, we avoid processing the battery power via two stages. Efficient and smooth control of the power drawn from the fuel cell and the high voltage battery is achieved by controlling the front end DC-DC converter in current mode.
48VDC / 400VDC, 40KHz PUSH PULL CONVERTER
Fuel Cell Input
48VDC
120/240V , 60 Hz
Iin C
T1 T
TR11:
L
VDC
C2 C
Lb Lb
S1 S
S3 S
L
C L
iA C5iB
A N B
120V/240VAC, 20KHz PWM INVERTER
AC Output
Vin
D^
D D^
D
N1 N
N2 N
IDC
Battery Backup
Vbatt Vbatt
Note: Components shown in dotted boxes are not considered for cost evaluation
3 3
N K1 K
iT1 I^ T2 Figure 1: Schematic of the TAMU Fuel Cell Inverter System
iD^
iL
iC
A
B
N
iAO iBO
Pin 010000. 95 0 W.^95 11050 W (1)
A nominal fuel cell input voltage, V (^) in= 48VDC, is assumed. Output voltage, V (^) o = 400VDC Designing for the low input line condition (V (^) in=42VDC), input current Iin from the fuel cell is,
I in 1105042 VW^ 263 A (2)
The push pull DC/DC converter shown in Figure 1 comprises of two switches, T+ and T-. At the maximum duty ratio of 0.45, rms current rating IT of the switches are,
I T Iin 0. 45 176 A (3)
IRFP260N (200V, 50A) MOSFETs with 4 devices in parallel in each leg are then chosen. High frequency transformer: For obtaining an output voltage of 400VDC for the push-pull converter, a turns ratio of K=5 is selected for the transformer. Center taps are available on both the primary and secondary sides as shown in Figure 1.
The VA rating of the transformer is defined as the sum of the total primary and secondary
winding VA divided by two,
VA (^) Tr 21 Vin 2 Iin 2 2 Vin K IKin 2 1. 5 Vin Iin 1. 5 42 263 16600 W 17. 0 kVA (4) Voltage ratings of the transformer are selected as, Primary voltage=80V, Secondary voltage=400V Diode ratings: The reverse blocking voltage is equal to the DC link voltage 400V. Since each diode is clamped to the mid-point of the DC-link (200V), each diode can be rated for 300V. The rms current through the diode, ID , is given by
I D KI in 2 37. 2 A (5)
Therefore, 60EPU04 (400V, 60A), fast recovery diodes are selected. Design of Current Mode PWM Controller: The DC-DC Converter uses the 3-terminal push-pull topology to boost the 48V from the fuel cell to 200V at a switching frequency of 40kHz. The push-pull DC-DC converter is controlled by means of a high speed PWM controller UC3825B (datasheet attached in Appendix C ). The special features of this controller are: suitability for current control; soft start; over current and under voltage protection; low propagation delay; high current dual outputs and low cost. Current mode control has numerous advantages over simple voltage mode control, including making the converter respond faster to load changes. In particular the UC3825B is suitable for the fuel cell inverter application because it allows direct control over the power drawn from the fuel cell. The error amplifier output in the outer voltage loop defines the level at which the primary current (in the inner current loop) will regulate the pulse width and output voltage. Pulse-by-pulse symmetry correction is a feature of current mode control and thus is essential for flux balancing the transformer in the push-pull topology. Design methodology for the current mode controller is as follows, Timing section:
Oscillator frequency=40kHz; period=25 s From the UC3825B data sheet, for a maximum duty cycle of 0.9, we have
C RDF nF
R mA VD k
T T MAX
T MAX
10 ( 13 ) 10 ( 13 0. 9 )^3
3 3
2
which yields a T (^) ON =22.5 s, T (^) OFF=2.5 s. Power input to the DC-DC converter, P (^) in is
Equivalent ramp downslope voltage V (^) SL available across the sense resistor is,
V SL '^ 7. 1 2. 5 10 ^3 0. 01775 V s (11)
Slope of the oscillator waveform V (^) OSC is,
VOSC 221.^.^85 V s 0. 08 V s (12)
If the amount of inductor downslope voltage to be added to the oscillator waveform is 75%, then a resistive divider with resistors 10k and 30k can be selected. Input Capacitor : Selecting a proper input capacitor C1 (Figure 1) contributes to the reduction in fuel cell input current ripple. In this section, the selection of C1 is detailed.
The average input current Iavg at full load is 263A.
Assuming a square wave input current, for a duty ratio of 0.9, the peak current I,
I 2630. 9 292 A (13)
and the RMS current Irms is,
I rms 292 0. 9 277 A (14)
Therefore the RMS capacitor current Ic,rms ,
I c , rms Irms^2 Iavg^2 92 A (15)
Based on the rated ripple current, 4 Rubycon Aluminum electrolytic capacitors 22000 F, 100V each are selected.
The simulation results for a 10kW load on the system are presented in Appendix B. Vds1, Vds2 are the drain to source voltage across the MOSFETs T1 and T2 respectively. V (^) DC is the output voltage.
3.2 Inverter Design Procedure for the 10kW TAMU Fuel Cell Inverter System
The schematic of the DC-AC Inverter circuit is shown in Figure 1. The inverter produces two single-phase outputs, Phase-A and Phase-B. It is comprised of two half bridge inverters each supplying a separate single-phase load at 120VAC, 60Hz. Consider the case when Phase-B is not loaded and Phase-A is supplying full load (5000VA). The peak amplitude of the fundamental frequency component is the product of ma and ½ VDC , where ma is the modulation index. A modulation index of 0.9 is assumed for this design.
The fundamental component of the inverter Phase-A output voltage V (^) AO is,
V AO , 1 ma V 2 DC sin( 1 t ) 0 ma 1 (16)
The switching function sw 1 of the half bridge inverter is
sw 1 0. 5 02.^9 sin 1 t higherfrequencyterms (17)
The Phase-A output current (i AO) is assumed to contain fundamental and third harmonic
current components due to presence of nonlinear load. The current i AO can be expressed as,
i AO 2 I 1 sin( 1 t 1 ) 3 I 3 sin( 3 1 t 3 ) ... (18)
The current through the IGBT, S1 (i (^) sA) is given by
- (^92) cos cos( 2 02. (^93) cos cos( 3 ) ...
22 sin( ) 23 sin(^3 ) ... 1 1 1 1 3 3 1 3
1 1 1 3 1 3
1
I t I t
I t I t
i (^) sA sw iAO (19)
Assuming the load current i (^) A to consist of only fundamental (I1) and third harmonic component (I3), we have,
I A , rms I 12 I 32 (20)
3.3 DC-AC Inverter Output Filter Design Procedure
Figure 3 shows the topology for the output L-C filter. A transfer function is developed from the schematic. The assumptions used in the analysis are, the output filter is lossless and the third current harmonic current is 70% of the fundamental current frequency.
Figure 3: Output Filter
The transfer function for this type of filter is described by the equation
,(^2 )
, ,
, L C Ln L C
C Ln in n on nX X jZ nX X
jX Z V
H V
Where
H n - transfer function
Vo , n - output voltage harmonic
Vi , n -^ input voltage harmonic
X C - capacitive component of impedance
X L - inductive component of impedance
Z L , n - impedance
n - harmonic order
For H 1 1 ; or X^ L XC , then
H 1 jZjXLC , 1 ZXL , C^1 1. (28)
j n X (^) L
- j X C n Z L 1 n
Vi,n (^) V o , n
At no load, Z L , 1 , therefore equation (27) is
C
L C L n C
X
n X X n X
H X (29)
In order to satisfy a THD requirement of less than 3%
2 1 1 0.^03 XX^34^ n.^3332 X n X CL C L ^
Non-Linear Load
An equivalent circuit used in finding filter characteristics for a non-linear load is shown in Figure 4. jhXL
-jXC h V (^) h I^ h
Figure 4: Equivalent Circuit for a Non-Linear Load The transfer function for this schematic is described by equation
Vh (^) XjhXC Lh 2 X (^) XCL I h.^ (31)
Where
V h - equivalent voltage
h - harmonic order
I h - current at h harmonic
X C - capacitive component of impedance
X L -inductive component of impedance
equation (31) can then be shown as
where f 1 60 Hz , the inductance will be L 123 H.
To find the capacitor impedance use the equation (30), to get X C 148. 9 , then using
C 2 f 1 X C
where
C - capacitance
X C - capacitor component of impedance
f 1 -fundamental frequency
and f 1 60 Hz , capacitance will be C 18 F.
Simulation results for a 10kW load on the system are presented in Appendix B.
4. 4 .0 0 ScSchheemmaattiiccss
The following detailed schematics are attached in Appendix A. A1. DC-DC Converter: complete design schematic A2. DC-DC Converter voltage feedback and protection circuit details A3. Inverter power circuit and gate control A4. Inverter voltage and current sensing and protection circuitry (Sheet 1) A5. Inverter voltage and current sensing and protection circuitry (Sheet 2) A6. DSP Control board schematic (Sheet 1) A7. DSP Control board schematic (Sheet 2)
5. 5 .0 0 BiBillll ooff MMaatteerriiaallss
In this section, a detailed bill of materials is developed for the DC-DC converter and DC- AC inverter subsystems. The components in the bill of materials are shown in schematics in Appendix A. Table 1: Bill of Materials for DC/DC Converter, Bulk Capacitors and its associated control & protection circuitry (refer Figures A1-A2 in Appendix A) Description Type Rating Quantity MOSFETs IRFP260N 200V, 50A 8 PWM Controller UC3825B 1 Opto-isolated gate driver
HCPL3120 2
Power Diodes 60EPU04 400V, 60A 4 Input Capacitor Electrolytic (^) 100V, 22000 F 4 Bulk Capacitors Electrolytic (^) 250V,4500 F 2 Transformer 17kVA, 400V,38Arms
Inductors Coupled (^300) H, 38A 2 Sense resistors 0.01ohm,75W 4 High frequency capacitor
Film (^) 1200V, 0.1 F 1 Snubber resistor 500ohm, 10W 2 Snubber capacitor 1000V, 150pF 1 Power resistors 56k, 7W 2 Power diode 600V,15A 1 DC Input connector 1 Control input connectors 6 Op-amp LF347 1 Op-amp LF356 1 3-input NOR gates CD4023 1 2-input NOR gates C4011 1 Thermal switch 5R13-90M 1 Power supply 48IMP12-051515-7 1 Heatsink 1 LEDs 5 Switches 2 LCD Display 1 Zener diodes 4 Resistors 2W 4 Resistors 1W 2 Resistors 0.25W 37 Potentiometers 10k 1 Potentiometers 2k 2 Capacitors 50V 17
Table 3: Bill of Materials for DSP Control Board (refer DSP Schematics in Appendix A) Description Type Quantity DSP TMS320LF2407 1 CMOS AND gate 74LCX08 1 Serial communication IC
Max232 1 Signal translator P15C3245 1 D/A converter TLV5619 1 Voltage regulator TPS7333 1 7.372MHz oscillator Xc263 1 Zener diode LM4040 1 Ferrite beads 3 Resistors 0.25W 11 Jumpers 5 Capacitors 19 RS232 header 1 Headers 1
6. 6 .0 0 CoCosstt EEvvaalluuaattiioonn
With the practical experience gained by the working budget, the team’s industry partners and the faculty advisors, the team was able to make well-informed design decisions to aggressively lower the cost of the final 10kW design and 1.5kW prototype. The TAMU fuel cell inverter team’s approach to reducing the cost of the inverter by reducing the number of high cost switching devices by adopting push-pull topology, using a low cost PWM DC-DC controller and including an efficient DSP DC-AC control board. By use of the push–pull topology the number of MOSFETs was minimized to half that needed by a full bridge topology. IGBT’s were reduced in the inverter by use of the half bridge topology as opposed to the full bridge topology. The analog PWM controller provided a low cost solution to control of the DC-DC converter. It provides a single chip control solution opposed to complex discrete analog hardware. DSP control of the DC-AC inverter provides sophisticated control at low cost. Further, the DSP enables software control of the inverter and adaptability for stand-alone and utility interface modes. Software control translates into efficiency in human capital reducing costs of analysis, troubleshooting, development and manufacturing of the fuel
cell inverter. The use of the DSP allows a seamless interface with other components of a power management system, saving integration time and human resources. The topology of the TAMU Fuel cell Inverter System employs a high voltage battery floating on the DC-link. This approach does not add any additional power processing cost for load management. The cost for the power components of the TAMU Fuel Cell Inverter system were calculated by developing the cost of the DC-DC converter and the DC-AC inverter and adding the two components together. The cost analysis was based on the schematic shown in Figure 1 and the 10kW design procedure detailed in this report. The results of the cost analysis for the DC-DC converter are seen on the normalized spreadsheet Table 4 and the results of the DC-AC inverter costs are seen in Table 5. As per the cost analysis spreadsheet provided by the 2001 Future Energy Challenge Committee, the cost of the DC-DC converter was $598.09. The cost of the DC-AC inverter $198.69. The total cost of the TAMU Fuel Cell System was $796.78. It should be noted that the cost analysis spread sheet (Tables 4 & 5) do not give the absolute cost and assumes a fixed cost for control and packaging. These costs are highly dependent on the type of design and the number of units manufactured per month. The TAMU inverter control is based on a low cost DSP (TMS320C24X). Our design and experimental prototype has demonstrated that sophisticated control algorithms can be implemented on this DSP platform. Appendix D details a press release from Texas Instruments and lists a cost of $2.98 for the TMS320C24X DSP employed in the TAMU inverter design. The TAMU Fuel cell Inverter Team believes that with a detailed analysis of the control circuit and the ancillary components, this design can be mass produced and marketed for an amount below the target cost of $500.