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with more than 100 circuits | | haviAugust 1977 - 'eiouble issue-90p contents É HF, RECEIVERS E o7 AUDIO, Music MISCELLANEOUS di o nais a corais Vital er os e mi 8 complementary tu selectiva tina 78 carhesaligi slam E o o 67º CMOS atom ciais 8 Re dE ipi el 104 DE e no 88 kaybgord decoder ss Ioudipenkor delay ui 81 keria handkaenia E mero pros, 78 LED ogie ita Ri redes cancel lr rr da Ra o pre OO o a a eia sia comer dinda a stereo ns io 18 apeed controle tor motora E qu o mae % e! ong re e” | 7 ondas z pa paira E! mn 60 UMA TIO corona 9 2 Wear tado om : E qua ss 15 W bridge ampli tor cai ao Rs Design iDeas PowER suppLiEs od 5 adaiaadi a n Afro rag level contro 6 Duromai Nic arg 13 capectancstrequanoy contei 111/11) 3 Tmvere 3 cos sos Sê rim a commontass vital tatha o mis 10% comp lg sam”. 08 e poi fem Bs co u complementar emitia ogir Ea Tra ge 15 Frequanoy ditar arg one TUN 30 two TUNvortag doublor ú fraqueney douta ui 4088 à Uvolao qm “o frequency-veltoe comvertor é O MOR E LG rasopencs mar” 108 SiS rogair | à Tool ie E noise cancalin rar 7 nominverting integrator .. . . 83 TESTING AND MEASURING paset-ris tc top E E fita - no Re uai 28 cubo oi probe. = self-shifting register . o 95 auto Ara a e rom a 7 capacitance frequency converter a 3 aiparsaar pg complementary twin-T solective filter 72 transistor solar cell . 100 né using LED efrento dos | E mona O voltage controle LED brighees: e q a Sono Opte ii 82 eight-channel multiplexer 73 voltage-frequency converter és BRs e eqund ot CAD a neo 105 Domesnie su É addon timer tor ancozesarmendiosiook .. 49 Phsemeer oi a CMOS alarm circuits . so spot-frequency sinewave generator . 25 o timebase input circuit 2 “rito voltage comparator & electronic westhercock 48 voltage-frequency converter a fado calo Rs É mute puros imo uh ! 5 net fo imo hear a ceu mermo % “alone la Us campo ge com E abro” eo ê per o 10 oie olotmonio ba FUN, GAMES, MODEL BUILDING rode speed conto! 7 pd gen O a restin s50 asa a es A io ortig 3a long interval timer 16 nv O The drawback of most analogue timers (monostable circuits) is that, in order to obtain reasonably long intervals, the RC time constant must be correspondingly large. This invariably means resistor values in excess of | M$), which can give timing errors. due to stray Icakage resistance in the circuit, or large electrolytic capacitors, which again can introduce timing errors due to their leakage resistance. The circuit given here achieves timing intervals up to 100 times longer than those obtainable with standard circuits. It does this by reducing the charging current of the capacitor by a factor of 100, thus increasing the charging time, without the need for high value charging resistors. The circuit operates as follows: when the stari/reset bution is pressed C1 is discharged and the output of ICI, which is cone: A Qd E NOu qu (ONO RR inverting input of comparator IC? is at a lower potential than the non-inverting input, so the output of IC2 goes high. The voltage across R4 is approximately 120mV, so Cl charges through R2 at a current 'of around 120nA, which is 100 times lower than could be” achieved if R2 were connected direct to positive supply. É) Of course, if CI were charged from à | constant 130 mV it would quickly reach this voltage and would cease to charge, However, the bottom end of R$ is retumed to the output of ICI, and as the voltage across CI zises so does the output voltage and hence the charging voltage applied to R2. When the output voltage has risen to about 7.5 volts it will exceed the voltage set on the non-inverting input of IC2 by R6 and R7, and the output of IC2 will go low. A small amount of positive feedback provided by R$ prevents any noise present on the output of ICI from being amplified by IC2 as it passes through the trigger point, as this could otherwise give rise to spurious outpt pulses. The timing interval js given by the equation T; RaCi (+ PEA PS In + a Da AI pia pit te component values give the inlrval 100-+ CI, where CI is in microfarads, e.g. if GL JS 1u the interval is 100 seconds, E is evident from the equation that the timing interval can be varied linearly by replacing R2 with a IM potentiometer, or logarith- mically by replacing R6 and R7 with, say, a 10 k potentiometer. voltage- frequency converter eloktor julylaugust 1977 Using only à few components and an inte- arated switching circuit it is possible to construct a highperformance voltage- frequeney converter. With the component values shown in the diagram, the conversion ratio has a linearity of approx. 15%. An input voltage from 0 V... 10V will produce a corresponding 0... 10kHz squarewave output voltage. By imeans of potentiometer Pi, the circuit can be adjusted so that an input voltage of O V will produce an output frequency of O Hz, The components which determine the fre- quency are resistors R2, R3, R5, PI and capacitor C2. Using the iormulae Shown in the diagram, the conversion ratio of the circuit can be altered so that the circuit can be used for a number of different applications. When calculating the product of T=1.1RsC, care should be taken to ensure that this is always less than half the minimum ouíput period, Le. the positive output pulse should always be at least as long as the negative pulse. RAYTHEON product specificarions 8 nov 0480. 1p6 sen OpmASea ten-nao capacitance- frequency converter G. Geiger This simple circuit will produce a pulse train whose (average) frequency is directly pro- portional to the value of a capacitor which is to be tested. This waveform can thus be fed direct to a frequency counter which will then indicate the capacitance value in (tens of) picofarads. The circuit is based on 4 monostable multi- vibrator whose pulse width is given by: t=Ck- RxcIn2. If Rxis fixed then t is obviously proportional 0 & The Output pulse of the monostable is used to gate pulses from a stable oscillator through to the input of a frequency counter The number of pulses allowed through to be counted is thus proportional to t which ds proportional to Cx The reference oscillator uses a normal -27 MHz (model control band) third overtone crystal, but in this circuit it oscillates at its fundamental of around 9MHz. The monostable ICI is triggered by gate pulses from the frequency counter and allows pulses from the oscillator through Ná to the counter input. For the circuit to func- tion correctly the counter gate period must be longer than the longest period of the monostable, which with the values shown is around 20 ms. To calibrate the circuit a capacitor of accurately known value is connected across the Cy terminals and the preset is adjusted until “the count indicated is equal to the capacitance value in tens of picofarads, e. a 10 n capacitor should give a count of 1000 (not 100001). Any 1% silver mica capacitor 17 of greater than 1000 pF can be used for this purpose. In order that the calibration of the circuit should not be affected by tem- perature Rxb should be a good-quality multi- tum cermet trimmer. With the values shown the circuit will measure capacitance values from 1000 pF to IF, The range may be extended to include higher values by reducing the value of Rx. electronic scarecrow elektor july/august 1977] This circuit produces a good imitation of machine-gun fire, an effect which should be sufficient to deter the most persistent of The system consists Of three squarewave oscillators using CMOS inverter, all running at different frequencies. Oscillator N1/N2 r ! Lo set E is gates oscillator N3/N4, which in turm gates oscillator N5/N6. The result is that this last oscillator produces intermittent bursts of pulses, which makes the sound very realístic. The output can be taken from the right- hand side of C4 and be fed to an audio amplifier and suitable (weatherproof) loud- speaker. The circuit can also be used to produce background noises for devotees of war games. For this purpose a mixing amplifier stage, TI, may be provided, and inputs from several different oscillators combined. The use of different values for the mixer input Ici=n1..N6=4049 resistors R7, RI3 and R14 means that the input signals will be amplified by different amounts, thus producing a more varied sound. PJ], P2 and P3 can be adjusted experimentaiy to give the most pleasing results, IS (4808054) q A; E 1] the counter a reset pulse. A manual reset button is also provided. Any other desired cycle time up to the previously mentioned maximum may also be accommo- dated, but obviously some counts will require more or less diodes in the AND gate, The switchon and switch-off times of the equipment to be controlled are also deter- mined in the same manner. The binary equivalents of the on and off times are calculated and the appropriate counter out- puts are connected to AND gate inputs BI to Bá for switch-on and CI to C4 for switch- off. At switchon monostable N2/N5 is triggered, which sets flip-flop FFI, turning on Ti to activate the relay. At switoh-oft monostable N3/N6 is triggered, which resets FFI. Manual controls are also provided, If elektor july/ougust 1977 | several circuits are to be controlled with EA mi E SS =P NT A E ie UP GG) different switchon and switch-off times then N2, N3, N5, N6, FF] and TI may be duplicated. The one disadvantage of this circuit is that initially it must be reset at the time that the timing cycle is required to start, ie. there is no time-setting facility, so in the event of a power failure it would be necessary to wait until the correct start time before resetting the circuit. For this reason it is best to make the start Of the timing sequence occur at à convenient moment, such às in the morning” or carly evening. To make the clock input of the counter less susceptible to interference pulses on the mains waveform it may be 4 good idea to precede it by a Schmitt-trigger using two CMOS NAND gates as described elsewhere in this issue. super zener This circuit is intended primarily to produce a stable reference voltage in battery operated equipment designed for minimum current consumption. Despite the fact that only imA flows through the zener the output voltage showed a fluctuation of less than 1 mY for supply voltage variations of 10 to 30 volts, The reference voltage from the zener is applied to the noninverting input of a 741 op-amp, and the output voltage is the zener voltage multiplied by the op-amp É Ro + Rs gaine Vo = Vz x go This approach has two advantages. Firstly, a low temperature coefficient zener (5.6 V) can be used to provide any desired reference voltage simply by altering the op-amp gain. Secondiy, since no significant current is “robbed' “from the zener by the opamp input, the zener need only be fed by a small current. So that the resistance of the zener does not affect the output voltage the zener current must be fairly constant. This is achieved by feeding the zener via RI from. 10.30 the output of the op-amp. The zener current is Vo Ve so RI should be chosen to give a 2ener chrrent of about 1 mA. The reference voltage obtained from the op-amp output can supply currents of up to 15 mA. One point to note when using this circuit is that the supply voltage must be at least 2 V greater than the output voltage of the circuit. thermostat elektor july/august 1977 Although primarily designed to keep the water in an aquarium at a constant tempera- ture, this circuit is also suitable for à number of other applications, The circuit described here represents only the control section of the thermostat. In addition a temperature sensor and a triac relay, which at periodic intervals supplies the eating element with voltage, are necessary to complete the thermostat proper. A simple temperature sensor is provided by the NTC sensor described elsewhere in this isue, or by the temperature-voltage converter in Elektor 5, July/August 1975. A suitable triac circuit which triggers at the zero-crossing point (ie. when the load voltage and current are small, thus preventing interference and contact wear) is the solid state triac relay described in Elektor 11, March 1976, or, the “solid state relay* described elsewhere in this issue, The thermostat functions as follows: the water temperature is measured by the sensor (the NTC or diode), which is fixed to the glass on the outside -of the aquarium by sulating tape. Since only three amplifiers are needed for the control circuit of the thermostat, the remaining amplifier can be used to construct the NTC sensor. The voltage supplied by this circuit is compared in À2 with the preset value of PI and the LDR, and then amplified by a factor of 10. The amplified voltage is then compared in Al with the triangular voltage produced in 3, The result is a squarewave output volt- age, which triggers the triac circuit for longer or shorter periods. The desired reference temperature can be set by means of PI. The circuit also contains a second input which is sensitive to light This has the effect of raising the reference voltage of the thermostat so that the aquarium is allowed to get warmer during the day. With the component values shown in the dingram, the increase in temperature (the size Of which depends on RI and R2) is approx. 2ºC. The LDR may also be omitted if required. The 15V supply is providing that it is properly smoothed it need not be stabilised. The current consumption for the circuit is 3 mA, which rises to 6mA when light falls upon the LDR, and to ISmA when the LED at the output lights up. A certain amount of attention should be paid to the safety of the circuit; for this reason the NTC is placed on the Gutside of the tank, and the triac relay should be fitted with am opto-isolator so that there is no direct electrical connection between the input and the mains, not critical, and ) simple transistor tester This simple circuit checks the functioning and measures the current gain (hFE) of PNP or NPN bipolar transistors. It operates by feeding a known constant current into the base of the transistor and measuring the colleetor current. Since the collector current of a non-saturated transistor is hFE times the base current (which is known) it is a simple matter to calculate the value of hpE, and in fact the meter which measures the collector current can be calibrated directly Since both PNP and NPN transistors must be tested, two constant current sources are required, to provide a négative base current for PNÊ transistors and a positive base current for NPN transistors. The voltage dropped across the LED causes a constant current to flow through the emitter resistor of the TUP and a corresponding constant collector current, which flows into the base of the NPN transistor under test. This current cam be set to 10 HÁ by connecting a 50 HA meter between points B and É and adjusting PI The lower LED and TUN constitute the negative current source. Here again, this may be set to IONA by connecting a micro- ammeter between the lower points B and E, and adjusting P2. When a transistor is plugged into the appropriate socket a current of 10 HA will thus flow into the base and a current of hFE times this will flow through the mili- ammeter. The full-scale deflection of the milliammeter depends on the maximum hFE to be measured. Since the collector current is hFE, times the base current (which is “OL MA) à reading of | mA corresponds to an hpp of 100, so if a 5 mA meter is to hand it can be calibrated in hpF values from O to 500, which should be adequate for most run- of-the-mill transistors, However, for testing C' versions of small-signal transistors, which can have gains up to 800, a IO mA meter calibrated O to 1000 could be used, or à/ | lower £s.d. meter shunted to read 8 mA and calibrated O to 800. Readers may have noticed that it is actually the emitter current of the PNP transistor that is measured, which is of course 1 + hFE, times the base current. However, since few transistors have gains less than 50 the worst error introduced by this is less than 2%, which is probably less than the error of the milliammeter. Note. The use of LEDs as reference diodes for constant current sources is dealt with elsewhere in this issue. A/D converter eloketor july/august 1977 This circuit will deliver a number of pulses which is directly proportional to a negative input voltage, The conversion takes place in two stages, With $1 in position 1 capacitor CI will charge until the voltage across it is Va + Vf + Vy, where Va is the voltage at the slider of PI, Vf is the forward voltage of DI and Vx is the (negative) input voltage. SL is then Switched to position 2, when this voltage will appear on the left-hand side of R$. C2 will charge from CI through Rá until the emitter voltage of T1 equals Vb + Vbe 71 when TI will conduct. Since Tl and T3 are connected in the familiar thyristor tetrode configuration TI and T2 both tum hard on and C? discharges rapidy to Vbe T]- C2 will then recharee from C1 until TI again conduets, Fach time C2 is Charged CI loses a charge ÀQ equal to Vp C2. This cycle repeats until the voltage on CI has fallen below Vhp + Vbe 71, when TI will no longer conduct. If PI and P2 are adjusted so that Va + VE = Vb + VbeT] then this obviously Occurs when the voltage on CI has fallen to Va + Vf, ie. the portion of the voltage on É1 due to the input voltage Vy has been dissipated. The number of pulses, appearing at the collector of T2 is thus proportional to this input voltage, Since CI loses a charge AQ for each pulse the total number of pulses is obviously given by =, uere Meto: N= Sê slhere Qx is the charge on CI due t Vx, which is CIVx us N= 8 Thus N= ci, £Vx With the component values given the conversion factor is about 100 pulses per volt, Source: Electronic Engineering, September 1976 automatic NiCad charger H. Knote es ts Baoceoo| É lt is not generally appreciated that, if Nickel-Cadmium batteries are subjected to prolonged overcharging from chargers of the constant current type, their life may be considerably reduced. The charger described here overcomes this problem by charging at a constant current but switching off the charger when the terminal voltage of the battery rises, which indicates a fully-charged condition. The basic circuit described is intended to charge a single 500 mAh “AA” cell at the recommended charge rate of around 50 mA, but it can easily be extended at little cost to charge more than one cell Power for the circuit is provided by a trans- former, bridge rectifier and 5 V IC regulator. The cell is charged by a constant current source TI which is controlled by a voltage comparator based on a TTL Schmitt trigger NI. While the cell is charging the terminal voltage remains at around 1.25 V, which is below the positive trigger threshold of NI The output of NI is thus high, the output of N2 is low and T1 receives a base bias voltage from the potential divider R4/R5. While the cell is being charged DJ is it When the cell approaches the fully-charged E f6p 5 8, Tal Ca E ê E state the terminal voltage rises to about 1.45 V, the positive trigger threshold of N1 is excetded and the output of N2 goes high, turning off T1, The cell ceases to charge and Di isextinguished As the positive trigger threshold of NI is about 1.7V and is subject to a certain tolerance, R3 and P1 are included to adjust it to 1.45 V. The negative trigger threshold of the Schmitt trigger is about 0.9 V, which is below the terminal voltage of even à fully- discharged cell, so connecting a discharged cell in circuit will not cause charging to begin automatically. For this reason a start button $1 is included which, when pressed. takes the input of NI low To charge à number of cells the portion of the circuit enclosed in the dotted box must be duplicated. This has the advantage that, unlike chargers in which cells are connected in series, cells in any state of discharge may be placed on the charger and cach will be individually charged to the correct level, The disadvantage is that batteries of cells cannot be charged. However, up to ten AA cells may be charged if the circuit is duplicated the appropriate number of times two-TUN voltage doubler | elektor julylaugust 1977 This little circuit will produce a DC output that is almost twice the supply voltage. A square-wave input is required of sufficient level to tum TJ fully on and off. When TI is conducting, C2 is charged to just under the supply voltage. When TI is cut off, T2 starts to conduet and raises the voltage at the negative end of C2 to just under the positive supply level. This implies that the voltage at the positive end of C2 is raised to almost twice the supply voltage, so that C3 will ultimately charge to this levei. The circuit is remarkably efficient: the current drawn from the main supply is only marginally greater than twice the out- put current. In the example shown here, the efficiency is approximately 90%, The value for RI depends on the amplitude of the square-wave input: TI will require à base current of 0.5... | mA. (RCA application note) 12vjaoma Eca 22v/20mA JUL model speed control | | Ea ati a) The disadvantage of most simple speed controllers for model trains or cars is that they simply supply the motor with a fixed voltage. Consequently the speed does not remain constant, since the model slows down when climbing gradients and speeds up when going downhill. With model trains the setting of the control knob to maintain a particular speed also varies with the load that the engine is pulling. The circuit described here eliminates this problem by monitoring the motor speed and keeping it constant for a given control setting, irrespective of load. The circuit will operate with most models which use a DC tivo components, the back e.m f. generated by the motor and the voltage dropped across the armature resistance. The back em. is proportional to the motor speed and so motor speed can be sensed by measuring it, but the problem is to separate the back e.m.£ from the resistance voltage. If an external resistor is connected in series with the motor then, since the same current flows through it and through the armature resistance, the voltage drop across the series resistor wil be proportional to the drop across the armature resistance. In fact if the two resistances are equal then the two voltages will be equal, and the voltage across the series resistor can be subiracted from the motor voltage, leaving only the back em.f. The circuit monitors the back em.f. and adjusts the motor current so that, for a given control setting, the back e.m.., and hence the motor speed, remains constant. To simplify the description of the circuit it is ássumed that PJ is set to its mid-position and that R3 is equal to the armature resistance of the motor. The motor voltage is the sum of the back e:m.f. Va and the voltage dropped across the internal resistance Vr. Since a voltage Vr is dropped across R3 the output voltage Vo equals Va 2 Vr- The voltage ato He inverting input OF ICI s Va + Vis and that at o Va 2Ve-Vi the non-inverting input is Vi + ati These tivo voltages are equal, +2Vr= Vi o Simplifvina tie aqualiopo gives Vaio “which means that the back em.f, is always kept equal to the control voltage, so the motor runs at constant speed for a given setting of the speed control PI. PZ is used to compensate for the fact that R3 may not be equal to the armature resistance, by varying the amount of positive feedback to the non- inverting input. To set up the circuit a model is run and P2 is adjusted until the speed just remains constant on gradients and with different loads. Tf P2 is turned too far towards PJ then the model will slow down, but if P2 is turned too far in the opposite direction then the model will actually go faster when climbing a gradient. If the controller is to be used with several different models then they must obviously all be fitted with similar motors, otherwise the circuit would require readjustment whenever a different model was used. The output transistor TI should be fitted with a heatsink of around 4ºC/watt. stereo | noise filter elektor july/august 1977 The signal-to-noise ratio of an FM broadcast received in stereo is considerably worse than that of the same broadcast received in mono. This is most noticeable on weak trans. missions, when switching over from stereo to mono will considerably reduce the noise level. This noise reduction occurs because the Ieft-channel noise is largely in anti-phase to the right-channel noise. Switching to mono sums the two channels and the anti- phase noise signals cancel By summing only the high-frequency com- ponents of the signal it is possible to elim- inate the annoying high-frequency noise without. destroying the stereo image since channel separation is still maintained at middie and low frequencies. Each channel of the circuit consists of a pair of emitter followers in cascade, with highpass filters comprising R3 to R7 and C3 to C5 that allow crosstalk to occur between the two channels above about &kHz when switch SI is closed. When S1 is open the two channeis are isolated, but resistors R9 to R11 maintain a DC level on C3 to CS so that syitehing olcks do not ocour when S1 is The stereo-mono crossover frequency can be increased by lowering the values of €3 to C5 or decreased by raising them. 404 tremolo Ow Tremolo is one of the most popular effects used im electronic music. The tremolo effect is produced by amplitude modulating the music signal with a low-frequeney signal of between 1 Hz and 10 Hz, The effect gives »warmth and richness to the otherwise flat” sound Of instruments such as electronic organs. The most pleasing effect is produced »when the modulation waveform is sinusoidal. The music signal is buffered by emitter- follower TI and is then fed into op-amp IC1, whose gain can be varied by means of PI nO | The output of ICI is fed to the diode modulator DI/D2, the output of which is buffered by a second emitter follower T2. The sinusoidal signal is generated by an oscillator built around IC2, whose frequency can be varied between 1 Hz and 10 Hz by P2. The output level, and hence the modulation depth, can be varied by P3. Switch SI, when closed, disables the oscillator, which allows the” music signal to pass unmodulated. thermal touch switch eloktor july/august 1977 This ingenious touch switch is operated, not by skin resistance or capacitance effects, but by heat. The circuit utilies the negative temperature coefficient of silicon diodes. The 741 functions as à comparator with positive feedback and, for the purpose of describing the circuit it is assumed that the voltage drop across each diode is initially the same and that the op-amp output voltage is low. Due to the potentiometer effect of R4 and R$ the voltage on the non-inverting input of the op-amp will be pulled slightly lower than that on the inverting input. 1f DI is heated by touching it, its forward voltage drop will decrease, and so will the voltage on the inverting input of the op-amp. When this voltage falls below that on the non-inverting input the op-amp output will £o high. Positive feedback via R$ and R4 will pull the non-inverting input still higher, so that even if DI is subsequently allowed to cool to the same temperature as D2 the output will remain high. The switch is returned to its original state by touching D2. This will cause the voltage at the non-inverting input to fall, and when it is less than that on the inverting input the output of the op-amp will go low. The output of the switch can be used to control other circuits, relays etc. To indicate the state of the switch two LEDs may be driven by the circuit. If a brighter display is required than can be driven direct from the op-amp output then a two transistor buffer may be used to drive LEDs at a higher current Before use the circuit must be nulled to com- pensate for differences in the diode forward voltage drops and for the offset voltage of the 741. This is done by closing switch SI, which inhibits the positive feedback loop, and by adjusting PI until the op-amp outpuí voltage is approximately half supply (4.5 V). The circuit may also be used as a differential thermostat in such applications as solar heating. In this case one diode is mounted on the solar panel and the other on the hot water tank. PI is adjusted so that the circuit will operate when the required temperature difference between the two is reached. This is usually when the solar panel is 25 to 30C hotter than the tank inverter ava 4a mt E nando rafa This inverter circuit can be used to power electric razors, stroboscopes and flash tubes, and small fluorescent lamps from a 12 volt car battery. In contrast to the usual feed- back oscillator type Of inverter, the oscil lator of this inverter is separate from the output stage, which allows easy adjustment of the oscillator frequency to suit different applications. The oscillator circuit consists of a 555 timer connected as an astable multivibrator. The inclusion of Di ensures that the duty-cycle of the squarewave output is maintained at about 50%. The output of the 555 drives the base of TI which switches current through one half of the primary of the transformer, T2 is driven from the collector of T1 and thus switches current through the other half of the transformer winding on opposite half eycles of the drive waveform Zener diodes Dá and DS protect TI and TÊ from any highevoltage spikes generated by the transtormer. The voltage applied to the transformer primary is stepped up and the required high output voltage appears across the secondary winding. Depending on the application the secondary voltage may or may not be rectified. Components The transformer is a standard mains trans- former with two identical secondary windings or a single, centre-tapped second- ary. This transformer is, of course, driven in teverse, ie, the secondary becomes the primary and the output is obtained from the primary (which is now the secondary), It must be borne in mind that, since the inverter produces a squarewave Output, the RMS secondary voltage and peak secondary voltage are identical, This affects the choice of transformer for different applications. The required secondary voltage of the mains transtormer s given by Us = SM x 12 where 12 V is the inverter supply voltage Ur js the normal mains primary volt- age of the transformer. Up is the desired peak secondary volt- age An electric razor requires 240 V* RMS = 240 V? peak, so if airansformer witha 240 V. | primary is used the secondary windings Should each be 12 V or a single 12-0-12 wind- ing. For vibrator type (non-rotary) razors the oscillator frequency should be 50-60 Hz, so the value of CI should be 330 n and PI should be adjusted accordingly. Rotary zazors are less critical of mains frequency. When operated from the normal mains supply, fluorescent lamps receive a peak supply” voltage of around 340 V, which enables them to strike reliably. The trans- former secondary voltage should be caleu- lated with this in mind, which means that secondary voltages of eisht or nine volts will be suitable. Fluorescent lamps can be operated with improved efficiency at frequencies greater 4 than 50 Hz, and the transformer will also be more efficient. Choosing a value of 56 n for CI the oscillator frequency may be set to around 250 Hz. At frequencies much higher than this iron losses make the trans- former less efficient The current rating of the transformer depends upon the load. For electric razors and small fluorescent tubes up to 8 W, 500 mA secondaries will be adequate. Higher output powers may be obtained by choosing a suitable transformer, replacing T1 and T2 by higher power types and reducing the value of R3 and R4 (minimum 120 9), To power strobes and flash tubes the output Ê must be rectified and used to charge a 4 reservoir capacitor, which should be of à type rated for high discharge currents. The bridge rectifier should be rated to suit the peak output voltage. * UK, only. Overseas readers substitute the | | appropriate local mains voltage. 0...10 V. supply J. Borgman tor july/august 1977 The 723 is an extremely useful voltage stabiliser. It can be used for a wide range of supply voltages, with one limitation: the output voltage cannot be set at less than 2 V. This limit is set by the builtin differential amplifier. This problem can be solved by adding an “outboard” differential amplifier, such as the 130, in the circuit given nere tne 145 15 usea as a fixed-voltage regulator, providing a 14 V supply for the 3130. At the same time, the reference voltage output of the 723 is fed (via PJ) to P2 to provide a variable reference input fo the 3130, The gain of this opamp is set (by means of R7 and R$) at x2, so its output voltage will be twice the voltage at the slider of P2, The opamp output is buffered by T1, so that the supply can deliver up to 300 mà at any output voltage from O... 10 V. T1 should, of course, be provided with af adequate heatsink The current-limiting circuit in the 723 is also used. Tts current-sense input is connec- ted across R6; the value of this resistor sets the maximum output current. 2 resistor will limit the output to 300 mA; using à 6.8 9 resistor will bring the maxi- mum output down to 100 mA. The unregulated DC input to the circuit should be 20 V. This can be derived from a 16 V transformer winding, bridge rectifier and 1000 4F electrolytic capacitor. Initial calibration is quite straightforward. Set P1 to its maximum value (the slider turned down towards point K) and turn P2 all the way up (to point K). Now adjust Pl until the output voltage is exactly 10 V. P2 can be provided with a linear 0... 10V scale. It is advisable to use a high-quality potentiometer here; a multiturn type will prove useful if precise and stable setting of the output voltage is required. Alternatively, the circuit shown in figure 2 can be used. In this case, the output voltage can be altered in 1 V steps by means of the 10-position switch, and P3 allows fine-adjustment within each 1 V step. .— º A “ ci BDI37)TIP31 6; | 1) é! R6& -=|-" R5 *see text f o p1 4k7(5k) 0..104 Ea [ça Ra A? E] in E = c4 cs R3 RB e — E fot E 100]. 10n ma c8| c9 ES ps au |10n 1 PE + + E bo 77059 1 o A9a b e d e f E) h 1 Í 1k 1k 1k HA mk He dk 1 He dk Tk 1k 1k Ren É) eo ——s os Sia r Ae 1 1 ] | a O RS; A o) - + i l 1 l ] Si | ] ed E | [ ES =) Oo— =) o 4 p3 470k (O) O; 77059 2 Q threshold box Vea Rap) 3 Using a single LM324 quad op-amp IC a versatile threshold comparator may be constructed. The circuit will sound an aarm andjor energise a relay whenever the input voltage falls outside preset levels. The circuit may be adjusted to respond to the three following conditions. a) Input voltage falls below a preset level. b) Input voltage rises above a preset level. e) Input voltage falls inside a range of volt- ages defined by upper and lower limits (window comparator). Op-amps Al and AZ function as compara- tors, When the voltage on the inverting input of AI falls below the preset reference voltage on the non-inverting input the output voltage will go high. When the volt- age on the noncnverting input of AZ rises above the preset voltage on the inverting window comparator inputs 1 and 2 should be linked. When one or both outputs go high T3 will be turned on, energising the relay. The astable multivibrator built around A3 will begin to oscillate and will feed a train of short pulses at about 1.5 kHz to the amplífier comprising Ti and T2, thus sounding the alarm. A4 provides a stable 6 volt reference for the comparators. Coarse adjustment of the reference voltages is provided by P2 and PS and fine adjustment by PI and P4, Hysteresis is provided, which is variable by means of P3 and P6, This is particularly useful if the input signals are noisy. The relay should have an operating voltage of 9 Vor less, but if it is less then a suitable resistor must be included in series with the relay to drop the excess voltage. The maximum input voltage to inputs 1 and 2.is 25 V and the input resistance is | MO, but the input voltage (and resistance) can be increased by raising the value of R2 and R7 sawtooth- cco elektor july/august 1977 6.45v 2 E bois, e Nr nz dam NTNZS',40n 066 Res E fem é jm oo Pesa? Ei This sawtooth waveform generator which is . built round a current controlled oscillator is distinguished by its large sweep range. It is suitable for use in electronic music appli- cations, and the narrow output pulse also enables' the circuit to be used as a pulse- CCO for sample hold cireuits. The circuit consists of a controllable cur- rent source (TI, T2), a trigger (NI, N2) and a switch (T3). As soon as the circuit is switched on capacitor CI is charged by the current source. When the voltage across CI reaches the threshold value of NI, T3 is tumed on via NI and N2, and CI is dis- charged, after which the whole cycle repeats itself. The sawtooth output signal which is buffcred by FET T4 has a peak-to-peak value of approx. 1.3 V. With the component values shown in the diagram, the frequency can be adjusted from approx. 5 to 500 kHz (with PI) Although a higher output frequency is possible, there is à corresponding deterio- tation in the waveform. With C] = 5nó and Ri=1k, the frequency range runs from 0.5... 500 kHz. In place of NANDs inverters may be used, speed controller for motors M. Junghans Electric motors up to around a quarter horsepower can often be picked up very cheaply (ex-domestic appliances). They are extremely useful for driving bench saws, drilling machines etc. A speed controller that will set and maintain a near-constant speed under varying load conditions is a useful accessory in this type of application. The motor speed is sensed by an optical commutator consisting of a disc with 15 holes or slots around its periphery. This interrupts a light beam falling on à photo- transistor, which tums T1 on and off. Pulses from the collector of T1 are used to trigger a monostable IC1 whose Q output is inte- grated by R6/C4 and R7/CS to provide a DC output level which is inversely proportional to motor speed, The desired motor speed is set by PI and the inaias eloktor july/august 1977 desired and actual motor speeds are com- pared by T2. T3 and Tá form a trigger pulse generator which produces a pulse to fire the triac once every half cycle of the mains waveform. T2 varies the collector current of T3 and hence the charging current of C6. This in tum controls the point in the cycle at which the trigger pulse occurs. If the motor speed should tend to rise then the voltage on C4 will fall and the base current of T2 will be reduced. The collector voltage of T2 will rise, thus reducing the current through T3, and the trigger pulse will oceur later, causing the motor to slow down. If the motor tends to slow down the reverse will occur. The voltage on C4 will rise and the collector voltage of T2 will fall, thus increasing the coliector current of T3 and causing the trigger pulse to occur earlier. The trigger stage (T2... T4) is driven from an unsmoothed ful-wave rectified supply (A). When Tá fires, it will discharge C6 rapidly and then remain conducting until the supply voltage approaches the next zero- crossing. The result is that C6 is always dis. charged at the start of each half-cycle, so the position of the trigger pulses relative to the Zero-crossings is determined solely by the current through T3. In other words, the trigger pulses are synchronised to the mais The type of triac required will obviously depend on the motor used. t should be rated for at least three times (1) the nominal mains voltage; the current will depend on the maximum motor current, and a reason- able rule-of-thumb is to divide the motor power rating by the máins voltage and multiply the result by two. As an example, a 500 W/245 V motor would require à triac The trigger pulse transformer (Trl) can be wound on a type AL250 potcore: primary 80 ras, secondary 40 turns, both 0.1 mm enamelled copper wire (42 5. ans ea may be possible to dispense with the “disc-with- holes”. If the LED and phototransistor are mounted inside the motor casing, the motor itself may sometimes be used to periodically interrupt (or reflect) the light beam.